International Journal of Energetica (IJECA)  

https://www.ijeca.info   

ISSN: 2543-3717 Volume 7. Issue 2. 2022                                                                                                           Page 52-63    

   

 This open access article is licensed under the CC BY-NC license (https://creativecommons.org/licenses/by-nc/4.0/)              Page 52 

  

 

Optimal Controller Design and Dynamic Performance Enhancement of 

High Step-up Non-Isolated DC-DC Converter for Electric Vehicle 

Charging Applications 
 

Rajanand Patnaik Narasipuram
1, 2*

, Narendra Kumar Muthukuri
1, 2

, Subbarao Mopidevi
2 

 

  
1
Eaton India Innovation Center, Vehicle Group, Pune, INDIA 

2 
Laboratory of Electric Vehicles-EV Lab, Department of Electrical & Electronics Engineering, Vignan’s Foundation for Science 

Technology and Research, INDIA 

 
*
Corresponding author E-mail: rajanand.ee@gmail.com 

 

Abstract – Ideally, traditional boost converters can achieve a high conversion ratio with a high-

duty cycle. But, in regular practice, due to low conversion efficiency, RR reverse-recovery, and 

EMI (electromagnetic interference) problems, the high voltage gain cannot be performed, whereas 

CIBC (coupled inductor-based converters) can achieve high voltage gain by re-adjusting the turn 

ratios. Even though the leakage inductor of the CI (coupled inductor) makes some problems like 

voltage spikes on the main connectivity switch, high power dissipation, and voltage pressure can 

be minimized by voltage clamp. In this paper, a non-isolated DC-DC converter with high voltage 

gain is demonstrated with 3 diodes, 3 capacitors, 1-inductor, and a coupled inductor. The main 

inductor is connected to the input to decrease the current ripple. The voltage stress at main switch 

S is shared by diode D1 and capacitor C1 and the main switch is turned ON under zero current, 

hence it turns to low switching losses. This paper proposes two controllers like proportional-

integral (PI) controller and fuzzy logic (FLC) for dc-dc converter. Furthermore, it demonstrates 

the operation, design, mathematical analysis, and performance of DC-DC converter using 

controllers for efficient operation of the system is performed using simulations in MATLAB 2012b. 

 

Keywords: EV, Electric Vehicles, proportional integral, PI, On-Board Charger, fuzzy logic, FL, 

DC-DC 

 

Received: 08/11/2022 – Revised 28/11/2022 – Accepted: 05/12/2022 

 

I. Introduction 

Generally, environmental problems are caused by using 

non-renewable energy resources like fossil fuels, coal 

energies, etc. So by using PV, wind, and Tidal energy the 

environmental problems are reduced. This energy's 

applied at the input side or Distribution generation (DG) 

systems. Due to the climatic effects the voltage gain will 

be reduced [1-3]. It's the major drawback of using 

renewable energy sources. Moreover, it's an effective 

method, by using PV panels the number of PV cells is 

connected in series so that output voltage across the PV 

panel will be improved. A dark effect cannot be obtained  

 

 

 

[4-5]. The main advantages are a large transmission ratio, 

excessive voltage gain, and small size [6-8]. Input current 

ripples are the main consideration by using photo voltaic 

and fuel cell applications. Moreover, similar advantages 

are getting by using an ideal converter but during the 

practical conditions due to less Transmission efficiency, 

reverse recovery, and EMI problems high voltage gain 

will not achieve [9-10], [28]. However, many converters 

with different new techniques have been introduced for 

getting high voltage gain and high transmission 

efficiency [11], [24].  



Rajanand Patniak. N  et al. / International Journal of Energetica (IJECA) Vol. 7, N°2, 2022, pp. 52-63 

 

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Switched capacitors [11-13] and voltage lift techniques 

[14-16] have been introduced for improving the voltage 

gain. High current ripples at the input side are the main 

disadvantage so it reduces the performance and 

efficiency of the converter. Recently there are many non-

isolated coupled inductors-based converter are presented 

with different voltage clamping circuits [19-25]. They 

have the merit of high voltage gain; recovery of the 

leakage inductor's energy and low switching voltage 

stress are the main features and have the main drawback 

of high input current ripple. This makes problems in 

tracking the maximum power point (MPP) of PV panels.  

The proposed converter Figure 1 has excessive voltage 

gain by maintaining the required turn ratio. Moreover, 

using a Leakage inductor across the coupled inductor 

[17-19] produces voltage spikes across the main switch 

so it leads to more power discharge. By providing 

voltage clamping circuits the power stored in the leakage 

inductor will be recovered. So, it’s very important to 

provide voltage clamping circuits across the main switch 

S. 

 
Figure. 1. Schematic of Non-Isolated DC-DC converter 

 

The inductor is connected in series to the input side for 

minimizing the ripple currents at the input side. The 

voltage stress across switch S is shared by diode D1 and 

capacitor C1 and the main switch is turned on under zero 

current switching and conduction losses also decreased.  

II. Operating principle and Analysis 

Operating principle and analysis, consider to 1) all 

capacitors C and inductors L & Lm are taken large 

beyond any ripple voltages and currents, 2) Only leakage 

inductance ILK  is considered. 

II.1. CCM Analysis of the non-isolated dc-dc 

Converter 

It consists of 5 time intervals in a single switching 

period. Figure. II to Figure. VII shows a flow of the 

current path in CCM mode. The NS current of the non-

isolated coupled inductor is: 

 

                                                             (1) 

 

There is no simultaneous change in iL & iLS, the study 

state performance is given below. 

Mode I [t0<t<t1]: S and D3 are in conduction. In this 

period iLk is increased & equal to iLm. This mode of 

operation ends when ILK equals ILM. 

The equation of   iLK, 

 

            (2) 

 

The operating period of this mode is very small, where 

the value of LK is less and VLK is high. L becomes 

magnetized due to V1, ending of this first mode, ID3 

becomes zero due to the secondary current of the 

inductor. Using KCL, IS can take as 

      (3) 

 
Figure 2.  First Mode of Interval [to-t1] 

 

Mode II [t1<t<t2]: In this period the switch is still ON. 

The iLK improved and became more than then iLM. The 

iLK equation is written as follows: 

 

,      (4) 

 

D2 ON due to NS current. Thus, the D2 current improved 

from zero. Due to NS current, the C3 is charged .C1 

energy is discharged through the coupled inductor and 

C2.  L is energized because of V1. This period ends when 

S is OFF Using KVL the equations can be written as: 

 

                                                           (5) 

                  (6) 

                  (7) 

 

Figure 3 shows the second mode of the interval [t1-t2].  

Where n is the turns ratio of coupled inductor = NS/NP 



Rajanand Patniak. N  et al. / International Journal of Energetica (IJECA) Vol. 7, N°2, 2022, pp. 52-63 

 

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Figure 3. Second mode of interval [t1-t2] 

 

 

Mode III [t2<t<t3]: switch S is OFF, iL current flows 

through D1 and makes it conduction. D1 and D2 currents 

are written as: 

                               (8) 

 

                                           (9)  

 

The iLk is de-energized continuously until its equals the 

iLm. At this time iD2 becomes zero. 

 iLK can be expressed as follows:  

 

              (10) 

 

According to the above equation, less value of 

leakage inductor and high –ve voltage causes a high 

current slope.  Operating time is also small in this mode. 

Due to the de-energizing of inductors L and V1 the C1 is 

charged. This mode is when D2 OFF. Figure 4 shows the 

Third mode of the interval [t2-t3] 

 
Figure 4. Third mode of the interval [t2-t3] 

 

Mode IV [t3<t<t4]: The C1 is charged by V1 and storage 

energy in L. the leakage inductor voltage becomes 

charged in this mode. Its voltage is written as follows: 

(11)

    

        Where d4TS is the time interval of this period. C0 is 

energized due to NS current. D1 current will be written as: 

 

       (12)                

       

 This period ends when D1 current becomes zero. 

However, the slope of iLm is less than iLK, according to 

equation (1), the slop iD3 is +ve. In this mode the VL, VLm, 

can be expressed as: 

 

                               (13)     

                              (14)       

  (15)           

  

Figure 5 shows the fourth mode of the interval [t3-

t4]. 

 
Figure 5. Fourth mode of the interval [t3-t4] 

 

Mode V [t4<t<t5]: D3 is still conducting in this period. 

Due to iD3, the C0 is energized. The iD3 is expressed as: 

 (16)            

       

The diode D3 and the iLK slope calculate on the iL, iLm 

slope. Eliminating the iL, iLm ripples take the slope of iD3 

and the iLK tends to zero.  Then VLk will become zero. As 

in this equation, zero currents are possible when S 

becomes conduction like the beginning mode. In ZCS 

condition the switch is off. From KVL the voltage 

equation is written as: 

             (17) 

      (18)           

      

 VLK2 & VLK4 calculated from the iLk slope in 

periods II &IV. C2 and C3 are in series with NS and NP, 

iLk and iLm avg. currents are zero, according to Amp-Sec 

balance law. Considering the balancing law in C1, C2 

&C3, and C0 it can be justified that the average value of 

the diode current is equal to the io. So, the following 

equation can be obtained from Figure 2, 

 (19) 



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 (20)

                              

So, the ripples of the L & Lm are also not considered in 

the Vgain calculation. The istress of the S and D1 can be 

getting as given below: 

                (21) 

By not considering the 3
rd   

mode using this equation, 

the 4
th 

time interval can be obtained as 

 

           ⇒     (22) 

     

Due to (4) and (19) equations the voltage VLk2 can 

be found as: 

              (23)                       

 

Where, fswitch and RL, respectively. Join Volt-Sec 

balance basis on the LK by neglecting modes 1&3, the 

following equation is obtained. 

 

- 

                                  (24)                      

    

Applying Volt-Sec balance basis on the inductors the 

output is: 

      (25)                        

       

                     (26) 

                  

  

  (27) 

       

M=   (28)

     

As the above expression depends on switching 

frequency, RL, and ILK value the voltage gain of this 

new converter is controlled. Whatever, it does not 

contain significant change on the Vgain. The change of Q 

on the voltage gain is shown in Fig IV and it's nearly 

low. For example, fS=30 KHz, LK=5uH, RL=200 & 

D=0.5, the Q factor value is about 0.96.  

The actual Vgain is 96% of the conventional Vgain. 

The Vgain of the new converter is   

                                               (29)  

 

Figure 6 shows the fifth mode of the interval [t5-t6] 

and Figure 7 presents the current waveforms for the new 

converter in CCM Mode.                    

 
Figure 6. Fifth mode of the interval [t5-t6] 

 

 
Figure 7. Current waveforms for new converter in CCM Mode 

 

B.    Boundary Condition Mode (BCM) 

 

In this mode, at ending of the switching period 

currents of L and Lm become equal. Equations can be 

written as follows 

                                                 

(30)                     



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(31) 

 

             (32) 

 

Operating CCM mode in the new converter, coupled 

inductor Lm should be in CCM. Since NS and NP of the 

coupled inductor are series to the C2 and C3 according to 

the Amp-Sec balance Law, avg iLK & iLM is zero. At the 

end of this switching period, D3 current reaches zero in 

CCM. In BCM, D3 becomes zero at end of the switching 

period. So the value of iL, iLK & iLM becomes zero. At the 

final end of this operation -the iLm value is nearly equal to 

IL-0.5ΔiL.  If the Lm is in CCM, its maximum value of 

current is to be semi of its ripple. It implies the Lm is in 

CCM.  

                       (33)                  

 

For operating the converter in CCM, the Lmin must 

be calculated. It operates in CCM when the normal value 

of inductor current is better than the semi of its ripple, 

using eqn (31) Lmin can be obtained as 

 

                                   (34) 

 

Figure 8 shows the current waveforms in BCM. 

 
Figure 8. Current waveforms in BCM 

III. Design of the switching components  

Voltage and current stress are obtained by 

selecting the appropriate semiconductors of the new 

converter, according to the working principle the stress 

voltages of diodes and switches are formulated as. 

  (35) 

 (36) 

 

The D2(peak) current is obtained from equation (19). 

The value of Speak and diode D1(peak) currents is as 

follows:  

 

 

          (37) 

 

Based on Fig. III and taking equation (37), the 

current value of  D3(peak) can be written as given below: 

 

 

    (38) 

 

By neglect, the ESR of the CO, the VO ripple of the 

new converter can be given as follow 

 

 

 (39) 

 

According to [1], [18], [28], by taking into 

consideration a certain ripple value, CO size is designed 

using (39). Then, by taking the ESR of the selected 

capacitor, the voltage ripple is less than the desired 

voltage ripple. The peak-peak voltage across the ESR of 

the Capacitor CO can be taken as  

 

 

 (40) 

 

IV. Modelling of PI Controller  

The enhanced PI controller is the commonly used 

control technique in electric vehicle applications. Figure 

9 demonstrates the block diagram of the enhanced PI 

control technique  



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Figure 9. Block diagram of proposed PI controller for converter 

 

The error signal e(s) is sent to the PI controller i.e Kp and 

output control signal Q(s) of the enhanced PI controller 

of the proposed system is: 

 

V. Modelling of FLC  

The development of FLC is easier and provides effective 

flexibility for the sudden change in output loads. In 

addition, it provides effective and smarter dynamic 

responses and is very reliable in operation. The FLC set 

rules are defined below: 

1. Gaussian Method. 

2. Sigmoidal Method. 

3. Trapezoidal Method. 

4. Π Method. 

5. Triangular Method. 

6. Bell Method etc. 

 

Figure 9a shows the Block diagram of the proposed FL 

controller for the converter. 

 
Figure 9a. Block diagram of proposed FL controller for converter 

 

5.1 Fuzzification 

 The process of defining the fuzzification matches 

the input data with the predefined conditions with set 

rules to examine how well the condition of each rule 

matches that particular input instance. It gives a 

mathematical way to demonstrate each input and output 

variable in traditional language. Figures 9b, 9c, and 9d 

show the fuzzy membership function of the error input 

variable with 7 linguistics.  

Figure 9b. Membership function for single input 

Figure 9c. Membership functions error for change in error 

 

Figure 9d. Membership function for control signal 

VI. Simulation Results  

The whole analysis is done in MATLAB/Simulink, and 

analysis takes part in three cases which gives an open 

loop of dc-dc converter in case I, PV fed dc-dc converter 

with PI and FL controllers in cases 2 and 3 respectively. 

Table 1 shows the specifications of the PV module and 

DC-DC converter. 

 

 

 



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Table 1. Circuit Parameters  

 

Specification Parameters 

PV Parameters 

VOC 27V 

ISC 3.8A 

Parasitic resistor (Ro) 1Ω 

Parasitic capacitor (Co) 0.1mϝ 

DC-DC Converter Parameters 

Vin 27V 

Vout 290V 

Capacitors 
c1, c2, c3, c4=47Μϝ 

c0 =180μϜ 

Inductors 320Μh 

Coupled-Inductor 
Lm=100μH 

n=2.1 

Switching frequency 30Khz 

 

CASE (I): Open-loop simulation of new non-isolated 

coupled inductor-based high step-up DC-DC converter 

 

Designed for 225 watts of power and 27 input voltages 

achieving a high dc voltage of nearly 300 volts having 

less duty cycle 0.6. The conventional boost converter can 

achieve the same output voltage, but it requires 0.94 duty 

cycle for 300 volts. Figure 10 shows the Simulink model 

of coupled inductor-based dc to dc converter 

 

 
Figure 10. Simulink model of coupled inductor-based dc to dc 

converter 

 

The output voltage response of the new converter is 

shown in Figure 10a. It’s clear that the output voltage is 

300V at 60% duty cycle operates 36% less than 

conventional converter.  
 

 
 

 
Figure 10a. Output voltage and output current response of coupled 

inductor-based DC-DC converter 
 

Figure 10b shows the inductor and Diode current 

responses of coupled inductor converter. 

 
Figure 10b. Inductor and Diode current responses of coupled inductor 

converter 

 

Figure 10c shows the capacitor voltage responses of 

coupled inductor converter. 
 



Rajanand Patniak. N  et al. / International Journal of Energetica (IJECA) Vol. 7, N°2, 2022, pp. 52-63 

 

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Figure 10c. Capacitor Voltage responses of coupled inductor converter 

 

Figure 10d shows the Switch voltage responses (Diodes 

& main switch) of coupled inductor converter. 

 

 
Figure 10d. Switch voltage responses (Diodes & main switch) of 

coupled inductor converter 

Figure 6f shows the comparison of the voltage gains 

and switch stresses of the main switch for coupled 

inductor-based converter and boost converter. From 

the graphs, it is identified that the gain values of the 

coupled inductor-based converter are more than the 

boost converter for the same duty cycles. Also, the 

switch stresses are less for the coupled inductor-

based converter compared to the boost converter. 

Thus, the coupled inductor-based converter is 

superior in operation to the boost converter and in 

further sections, the dynamic performance of this 

converter is verified for the load changes using the 

PI controller and Fuzzy logic controller. 
 

 

 

Figure 10e. Comparisons of voltage gains and switch stress of 

coupled inductor-based converter and boost converter 
 

 

 

 

 

 

 

 

 

 



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CASE (II): closed-loop control simulation of PI-

controlled coupled inductor based dc to dc converter 

 

Figure 11a gives the simulation diagram of the proposed 

converter at an input voltage is 27 V, load current varies 

from 0.35 to 0.72 A and Figure 11b shows the voltage 

and current responses of dc-dc converter using a PI 

controller. 

 

 
 

Figure  
11a. Simulink model of PI-controlled coupled inductor based dc to dc 

converter 

 

 

 

Figure 11b. Voltage and current responses of dc-dc converter using PI 

controller 

 

CASE (II):  closed-loop control simulation of fuzzy 

controlled coupled inductor-based dc to dc converter 

 

The input voltage is 27 V, the output voltage 290V, and 

the load current are from 0.35 A to 0.72A with the FL 

controller. 

Figure 12a gives the Simulink model of Fuzzy controlled 

coupled inductor based dc to dc Converter and Figure 

12b shows the output voltage and current responses of 

FLC coupled inductor Converter for a sudden load 

change at 0.2 sec from 112.5W to 225W 

 

 

Figure 12a. Simulink model of Fuzzy controlled coupled inductor based 

dc to dc Converter 

 

 

Figure 12b. Output voltage and current responses of FLC coupled 
inductor Converter for a sudden load change at 0.2 sec from 112.5W to 

225W 

VII. Comparison Study  

Comparison for both PI and FUZZY controlled coupled 

inductor based dc to dc Converter. Figure 13a presents 

the Output voltage responses of PI & Fuzzy controlled 

coupled inductor Converter for a sudden load change at 

0.2 sec from 112.5W to 225W 

 

 



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Figure 13a. Output voltage responses of PI & Fuzzy controlled coupled 

inductor Converter for a sudden load change at 0.2 sec from 112.5W to 
225W 

 

Figure 13b shows the Simulink model of the coupled 

inductor with fuzzy logic controller maintaining the stiff 

output voltage of 300V even under the changes in the 

input voltage from 27V to 15V.  

 
Figure 13b. Simulink diagram of a coupled inductor with voltage 

variations using a fuzzy logic controller 

 

Figure 13c shows the output voltage response of 300V 

for the converter with fuzzy controller under the changes 

in the input voltage from 27V to 15V at 0.2 sec. 
 

 
Figure 13c. Output voltage response for input voltage variations applied 

at 0.2 sec from 27V to 15V. 

 

Figure 13d shows the comparison for both PI and fuzzy 

controller when sudden input voltage variations. 

 
Figure 13d. Comparison for both PI and fuzzy controller when sudden 

input voltage variations 
 

Thus, the fuzzy logic controller works effectively for the 

range of voltage variations at the input side from 15V to 

27V and delivers a stiff voltage of 300V. 

 

.  (41) 

VIII. Conclusion  

A non-isolated DC-DC converter with high voltage gain 

is designed and simulated in MATLAB R2012b version. 

The same output voltage of 300V is achieved for a very 

smaller duty ratio of 0.65 when compared with the 

conventional boost converter. It comprises 3-diodes, 3-

capacitors, 1-inductor, and a coupled inductor are 

utilized. The pressures switch S is reduced by utilizing 

the clamping circuit formed with diode D1 and capacitor 

C1. The dynamic response of the converter is analyzed 

for the disturbances on the load side from 112.5W to 

225W at 0.2 sec and the output voltage is maintained 

constant by using the PI and FLC.  The PI-controlled 

converter settles to the desired voltage with a dip of 1.2V 

at the disturbance point (0.2sec), whereas the fuzzy logic-

controlled converter gives a stiff voltage of 300V without 

any dip. Also, the fuzzy logic controller is verified for the 

disturbances at the input side for a range of 15V to the 

27V input voltage.  

 

Declaration 

 The authors declare that they have no known financial or 

non-financial competing interests in any material 

discussed in this paper. 

 The authors declare that this article has not been published 

before and is not in the process of being published in any 

other journal. 

 The authors confirmed that the paper was free of 

plagiarism. 

 



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